外文翻譯---單相spwm逆變器并聯(lián)解耦控制策略

1、<p><b>　　附錄6 外文文獻(xiàn)</b></p><p>　　Decoupling Control Strategy for Single Phase SPWM Parallel Inverters</p><p>　　Shun-Gang Xu,Jian-Ping Xu,and Tai-Qiang Cao</p><p>　　1

2、. Introduction</p><p>　　Parallel operation of inverters is an efficient way to enhance the capacity and reliability of inverter systems. The key issue of parallel operation is the distribution of the load cu

3、rrent. In an inverter parallel system, the amplitudes and phases of output voltages of all inverters should strictly equal to each other to guarantee that each inverter shares the same load current. Otherwise, the curren

4、t circumfluence and overload of some inverters in the inverter parallel system may exist. The cur</p><p>　　There are various techniques for the control of inverter parallel operation. Among these techniques,

5、 central control and master-slave control are easy to implement and have good current-sharing performance. However, these two control strategies work at the cost of system reliability because of conjunction operation amo

6、ng inverters.</p><p>　　In instantaneous-current control of inverter parallel system, there is a current bus to share the current signal among inverters and the instantaneous circumfluence is used to regulate

7、 the output current, each inverter has good transient performance and the parallel system has good current sharing performance. However, its analog signal communication is easy to be disturbed and the signal isolation is

8、 complicated, which decrease the reliability of the parallel system. </p><p>　　Independent control without interconnection droops the output voltage and frequency of inverters, the link among inverters is on

9、ly via power lines. Thus fewer interconnections are needed and the reliability of inverter parallel systems is improved. Traditionally, this control strategy assumes the output impedance of inverters is mainly inductive

10、due to high inductive component of the line impedance and the large inductor filter. Thus active power-frequency droop and reactive power-voltage droop </p><p>　　In this paper, a decoupling control strategy

11、for inverter parallel systems is proposed. The active power and reactive power of inverters in a parallel system are calculate by their corresponding output voltage and output current, and the output power information is

12、 shared by controller area network(CAN)bus communication. Then the active and reactive power circumfluence of each inverter is calculated and applied to regulate its corresponding output voltage and output frequency by d

13、ecoupling of th</p><p>　　2. Analysis of Single Phase PWM Inverter</p><p>　　Dual closed-loop feedback control is usually adopted to control single phase inverters.Fig.1 shows a dual closed-loop f

14、eedback control scheme with an inductor-current inner loop and a capacitor voltage outer loop. The capacitor-voltage outer loop adopts proportion-integral control to regulate output voltage, where andare proportional co

15、efficient and integral coefficient, respectively. The inductor-current inner loop uses proportional control to enhance the transient response of the inverter, is</p><p>　　In Fig.1,the power stage includes a

16、full-bridge configuration and an L-C filter, is DC link voltage, to are power switches, L and C are filter inductor and capacitor,is a sinusoidal reference voltage signal of the inverter,is the sum of inductor equivalen

17、t series resistance, switch on-resistance, and connection-line resistance. According to nonlinear control and feedback linearization theory, open-loop averaged output voltage can be characterized by</p><p>&

18、lt;b>　　(1)</b></p><p>　　where means the average value of x over one switching cycle and u is the control variable, which can take the values 1,0,or-1,depending on the state of switches ,,and.For the

19、 dual closed-loop feedback control inverter shown in Fig.1,the controller can be characterized by</p><p><b>　　(2)</b></p><p>　　From (1) and (2) ,the dynamic characteristics of the cl

20、osed-loop output voltage can be expressed in Laplace domain as</p><p><b>　　(3)</b></p><p>　　The single phase dual closed-loop inverter can be modeled by two terminal equivalent circu

21、its as</p><p><b>　　(4)</b></p><p><b>　　(5)</b></p><p><b>　　(6)</b></p><p>　　Fig.1.Block diagram of Single phase dual closed-loop inv

22、erter.</p><p>　　Frequency (rad/sec)</p><p><b>　　(a)</b></p><p>　　Frequency (rad/sec)</p><p><b>　　(b)</b></p><p>　　Fig.2.Bode diagra

23、m of the voltage gain and the equivalent output</p><p>　　impedance of the dual closed-loop inverter:(a)magnitude vs.</p><p>　　frequency and(b)phase vs.frequency.</p><p>　　Fig.3.Inve

24、rter equivalent circuit.</p><p>　　where is the voltage gain andis the equivalent output impedance. The bode diagram ofandare shown in Fig.2. </p><p>　　From (6), we can know that the equivalent o

25、utput impedance is closely related to the parameters of the output filter and the feedback control parameters. Let R be the resistive component and X the inductive component of equivalent impedance Z(s).The inverter equi

26、valent circuit can be shown as Fig.3.When, , and,the relations between the impedance ratio and the control parameters,, andare shown in Fig.4.</p><p>　　Fig.4.Relations between the impedance ratio R/X and co

27、ntrol parameters:</p><p>　　(a)R/X vs.,(b)R/X vs.,and(c)R/X vs..</p><p>　　From Fig.4, the equivalent output impedance trends to be resistive when PI control parameterandare increasing, and trends

28、 to be inductive when PI control parameteris increasing. In the design of dual closed-loop single phase inverter, the PI control parameters must be chosen carefully as they affect both the transient characteristics of th

29、e inverter and the current sharing performance of the inverter parallel system.</p><p>　　3. Analysis of Inverter Parallel System</p><p>　　Based on above discussion, the equivalent circuit of inv

30、erter parallel system of two inverter modular can be given as Fig.5, whereis load voltage, andare the output voltage and equivalent output impedance of inverter 1,andare the output voltage and equivalent output impedance

31、 of inverter 2.In the inverter parallel system, the active output power and the reactive output power of the inverter 1 can be expressed as:</p><p><b>　　(7)</b></p><p>　　Due to small

32、 difference the phase of output voltage between individual inverters, we can assume that,and.Therefore, we have</p><p><b>　　(8)</b></p><p>　　Similarly for the inverter 2, we have<

33、/p><p><b>　　(9)</b></p><p>　　Fig.5.The equivalent circuit of the parallel system of two inverter modular.</p><p>　　Fig.6.Structure of parallel operation system.</p>

34、<p>　　From above analysis we can know that the active/reactive power is related to the amplitude and phase of voltage, and the influence of output voltage amplitude and phase on active and reactive power is closely

35、related to the inductive component and resistive component of the output impedance of the inverter. When resistive component is dominating, active power is mainly depended on the amplitude of output voltage, and reactive

36、 power is mainly depended on the phase of the output voltage, and vice </p><p>　　4. Control Design</p><p>　　Fig.6 shows the structure of inverter parallel system. The digital signal processor TM

37、S320F2812 is adopted in the proposed parallel system; the inverters decouple the active power and the reactive power circumfluence to regulate the amplitude and the phase of the sinusoidal reference voltage signal. Each

38、inverter adopts instantaneous voltage and instantaneous current dual closed-loop feedback control. The inverters can operate not only independently but also in parallel. The CAN bus transfers in</p><p>　　Fig

39、.7.Decoupling control strategy.</p><p>　　Fig.8.Experiment wave of inverter parallel system: (a)steady current wave,(b)current wave with a sudden increasing load, and (c)current wave with a sudden decreasing

40、load.</p><p>　　In the parallel operation system, the differences between the output active power and reactive power of individual inverter lead to the asymmetry of output current among the inverters. The rel

41、ation between the active/reactive power and output voltage amplitude/phase is given by (8).In the single phase SPWM inverter which adopts dual closed-loop feedback control, output voltage tracks the amplitude and phase o

42、f the sinusoidal reference voltage signal. Thus, the output active and reactive power of</p><p>　　In the inverter, the output voltage and output current are sampled by digital signal processor (DSP) for the

43、calculation of output active and reactive power. All of the inverters share the active and reactive power by the CAN bus, each inverter calculates its corresponding active power circumfluenceand reactive power circumflue

44、nce.These circumfluence signals are decoupled to regulate the amplitude and the phase of reference voltage signal as shown in Fig.7.Therefore, each inverter outputs the sa</p><p>　　5. Experiment Results</

45、p><p>　　Two 2 KVA inverters are used in our experiment. In the parallel system, the output filter inductance is 500μH,the filter capacitance is 10μF,the DC input voltage is 200 V DC, and the AC output voltage i

46、s 110 V with 50 Hz. 6N137 is used to isolate the signal between the inverters and the CAN bus, the baud rate of CAN bus is set to 1 Mbps. The closed-loop control, decoupling arithmetic and the SPWM control signal are rea

47、lized by TMS320F2812 digital signal processor. Experiment results of the inver</p><p>　　6. Conclusions</p><p>　　This paper proposes a decoupling control strategy for inverter parallel systems. T

48、heoretical analysis and experimental results verify the feasibility of the proposed control strategy. This control strategy has the following characteristics:1)inverters can work independently or in parallel;2)CAN bus is

49、 used for the inverter parallel system; 3)the inverter parallel system supports hot-swappable operation and has good reliability and expansibility.</p><p><b>　　中文譯文：</b></p><p>　　單相S

50、PWM逆變器并聯(lián)解耦控制策略</p><p>　　徐順剛，徐建平，曹太強(qiáng)</p><p><b>　　簡介</b></p><p>　　逆變器并聯(lián)運(yùn)行是一種有效提高逆變器系統(tǒng)的容量和可靠性的方式。并聯(lián)運(yùn)行的關(guān)鍵問題是負(fù)載電流的分布。在逆變器并聯(lián)系統(tǒng)中，所有逆變器輸出電壓的幅值和相位應(yīng)嚴(yán)格相等，以保證每個逆變器有相同的負(fù)載電流。否則，逆變器并聯(lián)系

51、統(tǒng)中的一些逆變器可能存在電流回流和超載。電流回流可能也降低了逆變器并聯(lián)系統(tǒng)的效率和可靠性。</p><p>　　有各種技術(shù)可以控制逆變器的并聯(lián)運(yùn)行。在這些技術(shù)中，中央控制和主從控制比較容易實現(xiàn)，而且具有良好的電流共享性能。然而，這兩種控制策略因為逆變器之間的配合操作降低了系統(tǒng)的可靠性。</p><p>　　在瞬時電流控制逆變器并聯(lián)系統(tǒng)，存在一條電流總線用來共享逆變器之間的電流信號，同時，瞬

52、時回流用于調(diào)節(jié)輸出電流，每個逆變器具有良好的瞬態(tài)性能而且并行系統(tǒng)具有良好的電流共享性能。然而，其模擬信號通信容易受到干擾而且信號隔離難以實現(xiàn)，這降低了并聯(lián)系統(tǒng)的可靠性。</p><p>　　沒有聯(lián)網(wǎng)的獨立控制拉低了逆變器輸出電壓和頻率，逆變器之間的聯(lián)系只能通過電源線。因此需要更少的互連，提高了逆變器并聯(lián)系統(tǒng)的可靠性。傳統(tǒng)上，這種控制策略假定逆變器輸出阻抗主要是由于高線路阻抗和大電感濾波電感元件電感。因此采用有功功

53、率頻率衰減和無功功率電壓下降策略。然而，這并不總是真正閉環(huán)輸出阻抗也取決于控制策略和線路阻抗主要是低壓電纜的電阻。因此，輸出有功/無功功率和頻率/輸出電壓幅值之間有耦合關(guān)系。傳統(tǒng)的獨立控制可能會導(dǎo)致逆變器并聯(lián)系統(tǒng)的不穩(wěn)定。</p><p>　　在本文中，采用了逆變器并聯(lián)系統(tǒng)的解耦控制策略。逆變器的有功功率和無功功率在并行系統(tǒng)中通過相應(yīng)的輸出電壓和輸出電流計算得出，輸出功率信息通過控制器區(qū)域網(wǎng)絡(luò)（CAN）總線通信共

54、享。然后，分別計算每個逆變器的有功和無功回流，通過電源回流解耦來調(diào)節(jié)相應(yīng)的輸出電壓和輸出頻率。因此，提出的的解耦控制策略克服了獨立控制的逆變器并聯(lián)控制系統(tǒng)不互通和瞬時電流控制的缺點。運(yùn)用這一策略的逆變器并聯(lián)系統(tǒng)可以更好地實現(xiàn)電流共享性能，穩(wěn)定性好，可靠性高</p><p>　　單相PWM逆變器的分析</p><p>　　通常采用雙閉環(huán)反饋控制策略來控制單相逆變器。圖1顯示了一個電感電流內(nèi)環(huán)

55、和一個電容器的電壓外環(huán)的雙閉環(huán)反饋控制系統(tǒng)。電容電壓外環(huán)采用比例積分控制來調(diào)節(jié)輸出電壓，其中，和分別是比例系數(shù)和積分系數(shù)。電感電流內(nèi)環(huán)采用比例控制，以提高逆變器的瞬態(tài)響應(yīng)，是一個比例系數(shù)。</p><p>　　功率級包括全橋配置電路和LC濾波器，時直流母線電壓，到是四個電源開關(guān)，L和C是濾波電感和電容，是正弦逆變器的參考電壓信號，是電感的等效串聯(lián)電阻，切換電阻，連接線電阻的總和。根據(jù)非線性控制和反饋線性化理論，開

56、環(huán)平均輸出電壓可表征為</p><p><b>　　(1)</b></p><p>　　其中，是一個開關(guān)周期中x的平均值，u是控制變量，可以根據(jù)開關(guān),,和的狀態(tài)取值為1,0或者-1。在圖1所示的雙閉環(huán)反饋控制逆變器中，控制器可以表征為</p><p><b>　　(2)</b></p><p>　　

57、由（1）和（2）知，閉環(huán)輸出電壓的動態(tài)特性經(jīng)拉普拉斯變換可表示為</p><p><b>　　(3)</b></p><p>　　可以通過兩個終端等效電路對單相雙閉環(huán)逆變器建模</p><p><b>　　(4)</b></p><p><b>　　(5)</b></p&

58、gt;<p><b>　　(6)</b></p><p>　　圖1 單相雙閉環(huán)逆變器的結(jié)構(gòu)框圖</p><p><b>　　頻率（弧度/秒）</b></p><p><b>　　（a）</b></p><p><b>　　頻率（弧度/秒）</b&g

59、t;</p><p><b>　　（b）</b></p><p>　　圖2 Bode圖和等效輸出電壓增益阻抗的雙閉環(huán)變頻器：</p><p>　　幅度與頻率（b）相位與頻率</p><p>　　圖3 逆變器等效電路</p><p>　　其中，表示電壓增益，表示等效輸出阻抗。和的波德圖如圖2所示。&

60、lt;/p><p>　　從公式（6）我們可以知道，等效輸出阻抗輸出濾波器和反饋控制參數(shù)的參數(shù)密切相關(guān)。令R是電阻元間件，X是等效阻抗為Z(s)的電感元件。逆變器等效電路如圖3所示。其中，, ，；阻抗比和控制參數(shù),,和之間的關(guān)系如圖4所示。</p><p>　　圖4阻抗比R/ X和控制參數(shù)之間的關(guān)系：</p><p>　　（a）與（B）與（c）與</p>&

61、lt;p>　　由圖4可知，隨著PI控制參數(shù)和的增加，等效輸出阻抗是增加的，隨著PI控制參數(shù)的增加，等效輸出阻抗隨之減小。在設(shè)計雙閉環(huán)單相逆變器時，PI控制參數(shù)必須慎重選擇，因為它們影響將會逆變器的瞬態(tài)特性和逆變器并聯(lián)系統(tǒng)的電流共享性能。</p><p>　　3．逆變器并聯(lián)系統(tǒng)的分析</p><p>　　基于上述討論，雙逆變器的模塊化逆變器并聯(lián)系統(tǒng)的等效電路如圖5所示，其中，是負(fù)載電壓

62、，和表示輸出電壓和逆變器1的等效輸出阻抗，和表示輸出電壓和逆變器1的等效輸出阻抗。在逆變器并聯(lián)系統(tǒng)中，逆變器1的有功輸出功率和無功輸出功率可以表示為：</p><p><b>　　(7)</b></p><p>　　由于獨立逆變器輸出電壓的相位上的微小的差異，我們可以假設(shè),，，因此，我們得到</p><p><b>　　(8)<

63、/b></p><p>　　對逆變器2進(jìn)行類似處理，我們得到</p><p><b>　　(9)</b></p><p>　　圖5 兩個逆變器的模塊化并聯(lián)系統(tǒng)的等效電路</p><p>　　圖6 并聯(lián)運(yùn)行系統(tǒng)的結(jié)構(gòu)</p><p>　　從以上分析我們可以知道，有功/無功功率與電壓的振幅和相位

64、的有關(guān)，輸出電壓的幅值和相位及有功和無功功率與感應(yīng)組件和逆變器的輸出阻抗的電阻分量密切相關(guān)。當(dāng)電阻元件起主導(dǎo)作用時，有功功率主要取決于輸出電壓的幅度，無功功率主要取決于輸出電壓的相位，反之亦然。</p><p><b>　　4．控制系統(tǒng)設(shè)計</b></p><p>　　圖6為逆變器并聯(lián)系統(tǒng)的結(jié)構(gòu)。并行系統(tǒng)采用數(shù)字信號處理器TMS320F2812，逆變器解耦有功功率和的

65、無功功率回流來調(diào)節(jié)正弦參考電壓信號的振幅和相位。每個逆變器采用瞬時電壓及瞬間電流雙閉環(huán)反饋控制。逆變器不僅可以獨立操作而且可以平行運(yùn)行。 CAN總線傳輸逆變器之間的有功功率和無功功率的信息。</p><p><b>　　圖7 解耦控制策略</b></p><p>　　圖8逆變器并聯(lián)系統(tǒng)的實驗波形：（a）穩(wěn)定的電流波，（b）突然增加負(fù)荷的電流波形（c）突然降低負(fù)荷的電流

66、波形。</p><p>　　在并聯(lián)運(yùn)行系統(tǒng)，單個逆變器輸出有功功率和無功之間的差異，導(dǎo)致逆變器輸出電流之間的不對稱性。圖（8）給出的有功/無功功率和輸出電壓的振幅/相位之間的關(guān)系。在采用雙閉環(huán)反饋控制的單相SPWM逆變器中，輸出電壓跟蹤正弦參考電壓信號的幅度和相位。因此，逆變器輸出的有功和無功功率是可以由參考電壓信號的相位和振幅控制的。如果輸出的有功和無功功率與其并行系統(tǒng)中相等，則逆變器可以很好地分擔(dān)負(fù)載電流。&

67、lt;/p><p>　　在逆變器中，通過數(shù)字信號處理器（DSP）對輸出電壓和輸出電流進(jìn)行采樣進(jìn)行輸出的有功和無功功率的計算。所有逆變器通過CAN總線共享有功和無功功率，每個逆變器計算其相應(yīng)的有功功率環(huán)流和無功功率環(huán)流，這些環(huán)流信號通過解耦來調(diào)節(jié)參考電壓信號的幅度和相位，如圖7所示。因此，每個逆變器輸出相同的有功功率和無功功率，逆變器可以共享系統(tǒng)中的負(fù)載電流。</p><p><b>

68、　　5. 實驗結(jié)果</b></p><p>　　在我們的實驗中使用兩個2千伏安逆變器。在并行系統(tǒng)中，輸出濾波電感為500μH，濾波電容為10μF，直流輸入電壓為200 ??V直流，交流輸出電壓是110伏50 Hz. 6N137用于隔離逆變器和CAN總線之間的信號，CAN總線波特率設(shè)置為1 Mbps。 TMS320F2812數(shù)字信號處理器實現(xiàn)閉環(huán)控制，解耦算法和SPWM控制信號。逆變器并聯(lián)系統(tǒng)的實驗結(jié)果

69、如圖8所示，在穩(wěn)定狀態(tài)下，兩個逆變器分享電流非常好，在瞬態(tài)期間負(fù)載突然變化的情況下下，逆變器并聯(lián)系統(tǒng)仍然可以很好地工作。這表明，這兩個逆變器之間實現(xiàn)良好的負(fù)載分擔(dān)。</p><p><b>　　6.結(jié)論</b></p><p>　　本文提出了一種用于逆變器并聯(lián)系統(tǒng)的解耦控制策略。理論分析和實驗結(jié)果驗證了所提出的控制策略的可行性。這種控制策略具有以下特點：1）逆變器可獨